1. Field of the Invention
The present invention relates to switching power supplies, and more particularly to DC-to-DC voltage converters with inductive storing. Such converters, which transform DC voltages, have many industrial applications, especially in aeronautics, where they are used to generate, from the DC power supply of an aircraft, voltages of 5 volts, .+-.12 volts, .+-.15 volts, and so on, to supply various electronic devices.
2. Discussion of the Related Art
Generally, the operation of inductive storing converters is based on energy transfer cycles including a period of accumulation of magnetic energy in an inductive device, through a primary circuit, followed by a period of restitution of this energy in a load, through a secondary circuit. A converter whose inductive element is a single winding inductance is referred to as a "buck-boost" converter, and a converter whose inductive element is a transformer including at least two windings is referred to as a "fly-back" converter.
In this field the present invention more particularly relates to bidirectional voltage converters, which can transform energy both from the primary to the secondary and from the secondary to the primary. Such bidirectional converters are particularly adapted to supply complex loads (capacitive and/or inductive), accumulators or reversible devices such as electric motors, liable to feed back energy to the converter.
Exemplary bidirectional converters with inductive storing are described in U.S. Pat. No. 3,986,097 relative to fly-back converters, and in U.S. Pat. No. 4,736,151 and European patent application 336,725 relative to buck-boost converters.
A general object of the present invention is to improve bidirectional converters, whose operation will be first described.
FIG. 1a is a basic diagram of a conventional bidirectional buck-boost converter. Such a converter includes, on both sides of an inductance L, a primary circuit and a secondary circuit. The primary circuit includes a diode Dp and a chopping switch Tp that are connected in parallel, and interposed between a voltage source Vin and the inductance L. Similarly, the secondary circuit includes a diode Ds and a chopping switch Ts interposed between the inductance L and an output capacitor Cout. Capacitor Cout ensures smoothing of the converter output voltage Vout applied to a load Z. In practice, the switches Tp, Ts are electronic switches such as MOS or bipolar transistors.
As mentioned above, an operating cycle of the converter includes two periods, i.e. a first accumulation or storing period, having a duration Ton, during which a current Ip flows in the primary, and a second restitution or restoring period, having a duration Toff, during which a current Is flows in the secondary. FIG. 2 represents the total current I flowing through the inductance L during a cycle. In bidirectional operation mode, the storing period Ton in fact has a first phase Ton1 for restoring in the voltage source Vin an excess energy stored in inductance L during a previous cycle, followed by a second phase Ton2 for the effective storing of a magnetic energy in inductance L. FIGS. 1b and 1c are equivalent circuit diagrams of the converter during phases Ton1 and Ton2. During the phase Ton1, the inductance L generates the current Ip which is negative (diode Dp conductive--FIG. 1b). When this energy has been entirely restored, the storing phase Ton2 begins, where Ip is positive (Tp on, diode Dp blocked--FIG. 1c). Similarly, the so-called restoring phase Toff includes a first phase Toff1 for effectively restoring in capacitor Cout and in load Z the energy stored by inductance L during the phase Ton2, followed by a phase Toff2 of storing in inductance L an energy in excess provided to the capacitor Cout or to the load Z during the phase Toff1. FIGS. 1d and 1e are equivalent circuit diagrams of the converter during the phases Toff1, Toff2. It can be noted that the secondary current Is is positive during the phase Toff1 (diode Ds conductive--FIG. 1d), then negative during Toff2 (Ts on, diode Ds blocked--FIG. 1e), the load Z or the capacitor Cout behaving like a voltage generator. The energy stored during Toff2 is transmitted to the voltage source Vin during the phase Ton1 of the next cycle, which is a characteristic of a bidirectional operation.
A drawback of such a converter is that its efficiency decreases when the operation frequency increases, while, conversely, it is more advantageous to select high operation frequencies, ranging from 100 kHz to 1 MHz, to reduce the size and the bulkiness of the converter.
It is well known that the decrease of the efficiency with the increase of the operation frequency is especially caused by loss of energy in the switches during the switching periods. It will be noted that the energy lost in a switch during switching is equal to the product of the voltage across the switch by the current flow and the switching time. Switching-on losses and switching-off losses should be distinguished. In a converter operating in a bidirectional mode as represented in FIG. 2, the problem of switching-on losses of switches Tp, Ts is theoretically solved provided that each switching on is preceded by a conduction period of diodes Dp, Ds ensuring a switching-off voltage close to 0 (diode voltage). In contrast, each switching-off of a switch Tp, Ts causes the inductive element to react whereby the voltage across the switch increases abruptly, which in turn increases the switching-off duration through Miller effect. This phenomenon causes a loss of energy in the switches that is all the more significant as it occurs many times per second when the frequency is high. In addition, the abrupt increase in the voltage at the switching off generates spurious electromagnetic radiation. The same drawbacks occur in a fly-back transformer converter.
To avoid this drawback, one has proposed low-loss converters, in which the voltage edges occurring at the switching-off of the primary and secondary switches are smoothed by the addition of so-called "smooth-switching" capacitors.
FIG. 3 represents a low-loss fly-back converter 10. The converter 10 is, for example, of the type represented in FIG. 15 of European patent application 336,725. The converter 10 differs from the converter of FIG. 1 by the addition of two smooth switching capacitors Cp, Cs and by a specific operation mode including two transition periods which will be described hereinafter. Capacitor Cp is connected in parallel with the diode/switch circuit Dp/Tp of the primary and capacitor Cs is connected in parallel with the diode/switch circuit Ds/Ts of the secondary. In addition, since the converter is of the fly-back type, the inductance L of FIG. 1 is replaced with a transformer 1 which includes a primary inductance Lp formed by a winding of Np turns, and a secondary inductance Ls formed by a winding of Ns turns.
The operation of the converter is illustrated in FIG. 4. FIGS. 4a and 4b represent control signals Hp and Hs applied to switches Tp and Ts, respectively. FIGS. 4c and 4d represent currents Ip and Is respectively flowing through the primary Lp and the secondary Ls inductances of the transformer. FIGS. 4e and 4f represent voltages VTp and VTs across switches Tp and Ts. FIGS. 4g and 4h represent charge and discharge currents Icp and Ics of the capacitors Cp and Cs. FIGS. 4i and 4j represent spurious currents Iop and Ios, which are added to the primary Ip and secondary Is currents represented in FIGS. 4c and 4d.
As shown in FIG. 4, each operation cycle of converter 10 includes four distinct periods designated by their respective time durations T1, T2, T3 and T4.
Periods T1 and T3 are similar to periods Ton and Toff described above, except that the respective currents of the primary Ip and of the secondary Is flow through distinct windings Lp and Ls. Thus, period T1 comprises a restoring period at the primary where current Ip is negative (FIG. 4c--diode Dp conductive, Tp on or off), followed by a storing phase where current Ip is positive (diode Dp blocked, Tp on, Hp=1). Conversely, period T3 first includes a restoring phase at the secondary where current Is is positive (diode Ds conductive, Ts on or off) followed by a storing phase where current Is is negative (FIG. 4d--diode Ds blocked, Ts on, Hs=1). As represented by dotted lines in FIGS. 4a, 4b (during periods T1 and T3) there is an operation margin to turn on switches Tp and Ts, as long as diodes Dp and Ds are conductive.
The periods T2 and T4 are transition periods of short duration during which Tp and Ts are maintained off. In FIG. 4, T2 and T4 are not drawn to scale and are in fact of the order of 1/10 or 1/100 of T1 and T3. During these transition periods, capacitor Cp is discharged and capacitor Cs is charged, and conversely (FIGS. 4g and 4h). Diodes Dp, Ds are blocked, the energy stored in the transformer is not conveyed to the primary nor to the secondary.
Those skilled in the art will notice that the smooth switching capacitors Cp and Cs should not be mistaken for capacitors existing in so-called resonance converters, whose operation principle cannot be compared with the operation of the related inductive storing converters. Here, capacitors Cp and Cs are not means for transferring energy through the converter. On the contrary, capacitor Cp, Cs are selected so as to have charging and discharging periods T2 and T4 that are short with respect to periods T1 and T3 during which the energy is transferred within the converter.
The advantage of adding transition periods T2 and T4 and to use smooth switching capacitors Cp and CS is that, at the switching off of a switch Tp or Ts, the capacitor Cp or Cs associated with the switch is progressively charged and prevents voltage VTp or VTs from abruptly rising. As represented in FIGS. 4e and 4f, voltage VTp or VTs of the switch increases during the transition period T2 or T4 until it reaches its maximum value, which is equal to Vin+Vout*Np/Ns for VTp, and Vout+Vin*Ns/Np for VTs. The switching-off losses due to Miller effect are eliminated, or at least significantly reduced.
However the above converters have other drawbacks that will be described hereinafter.
Drawbacks caused by the use of smooth switching capacitors
The applicant has first noticed that despite smoothing of the rising/falling edges of VTp and VTs, the switchings still generate spurious electromagnetic radiation. More particularly, it has been noticed that these radiation are generated by spurious currents Iop, Ios due to a spurious oscillation phenomenon between the smooth switching capacitors Cp, Cs and spurious inductances present in the converter, which can be represented as a first inductance lp in series with Lp and a second inductance ls in series with Ls (FIG. 3). The currents Iop and Ios, represented in FIGS. 4i and 4j, occur after each switching of switches Tp, Ts, follow an oscillating state which slowly damps, and have a peak intensity which can be equal to the maximum values reached by Ip and Is at the end of T1 and T3. The current Iop flows in the whole loop formed by the primary circuit, including inductance Lp, the diode/switch/capacitor circuit Dp/Tp/Cp and the voltage source Vin, as well as the electrical connections between these elements. Similarly, the current Ios flows in the whole loop formed by the secondary. Thus, each primary or secondary loop generates an electromagnetic radiation due to an antenna effect, proportional to a magnetic flux .PHI.=S*.beta., where .beta. is the magnetic field generated by the spurious current and S is the surface area of the loop.